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Final Electrical Specifications
LTC1872 Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23
July 2000
FEATURES
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DESCRIPTION
The LTC(R)1872 is a constant frequency current mode stepup DC/DC controller providing excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the LTC1872 when the input voltage falls below 2.0V. The LTC1872 boasts a 2.5% output voltage accuracy and consumes only 270A of quiescent current. For applications where efficiency is a prime consideration, the LTC1872 is configured for Burst Mode operation, which enhances efficiency at low output current. In shutdown, the device draws a mere 8A. The high 550kHz constant operating frequency allows the use of a small external inductor. The LTC1872 is available in a small footprint 6-lead SOT-23.
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation.
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High Efficiency: Up to 90% High Output Currents Easily Achieved Wide VIN Range: 2.5V to 9.8V VOUT Limited Only by External Components Constant Frequency 550kHz Operation Burst ModeTM Operation at Light Load Current Mode Operation for Excellent Line and Load Transient Response Low Quiescent Current: 270A Shutdown Mode Draws Only 8A Supply Current 2.5% Reference Accuracy Tiny 6-Lead SOT-23 Package
APPLICATIONS
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Lithium-Ion-Powered Applications Cellular Telephones Wireless Modems Portable Computers Scanners
TYPICAL APPLICATION
R1 0.03 1 10k 220pF ITH/RUN LTC1872 2 3 GND VFB SENSE - NGATE 4 6 M1 D1 VIN 5 L1 4.7H C1 10F 10V VIN 3.3V
100 95
VOUT 5V 1A 412k
90
EFFICIENCY (%)
+
C2 2x 47F 16V
85 80 75 70 65 60 1 10 100 LOAD CURRENT (mA) 1000
1872 TA01b
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSE476M016R0150 D1: MOTOROLA MBRS340T3 L1: COILTRONICS UP1B-4R7 M1: Si9804DV R1: DALE 0.25W
100pF
78.7k
1872 TA01
Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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Efficiency vs Load Current
VIN = 3.3V VOUT = 5V
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LTC1872
ABSOLUTE MAXIMUM RATINGS
(Note 1)
PACKAGE/ORDER INFORMATION
TOP VIEW ITH/RUN 1 GND 2 VFB 3 6 NGATE 5 VIN 4 SENSE -
Input Supply Voltage (VIN).........................- 0.3V to 10V SENSE -, NGATE Voltages ............ - 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ..............................- 0.3V to 2.4V NGATE Peak Output Current (< 10s) ....................... 1A Storage Ambient Temperature Range ... - 65C to 150C Operating Temperature Range (Note 2) .. - 40C to 85C Junction Temperature (Note 3) ............................. 150C Lead Temperature (Soldering, 10 sec).................. 300C
ORDER PART NUMBER LTC1872ES6 S6 PART MARKING LTMK
S6 PACKAGE 6-LEAD PLASTIC SOT-23
TJMAX = 150C, JA = 230C/ W
Consult factory for Industrial and Military grade parts.
The q denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER Input DC Supply Current Normal Operation Sleep Mode Shutdown UVLO Undervoltage Lockout Threshold Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage VFB Input Current Oscillator Frequency Gate Drive Rise Time Gate Drive Fall Time Maximum Current Sense Voltage VITH/RUN = 0V 0C to 70C(Note 5) - 40C to 85C(Note 5) (Note 5) VFB = 0.8V CLOAD = 3000pF CLOAD = 3000pF VIN - VSENSE - 500
q q
ELECTRICAL CHARACTERISTICS
CONDITIONS Typicals at VIN = 4.2V (Note 4) 2.4V VIN 9.8V 2.4V VIN 9.8V 2.4V VIN 9.8V, VITH/RUN = 0V VIN < UVLO Threshold VIN Falling VIN Rising
q q
MIN
TYP 270 230 8 6
MAX 420 370 22 10 2.35 2.5 0.55 0.85 0.820 0.830 50 650
UNITS A A A A V V V A V V nA kHz ns ns mV
1.55 1.85 0.15 0.25 0.780 0.770
2.0 2.3 0.35 0.5 0.800 0.800 10 550 40 40 120
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC1872E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the - 40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD * JAC/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC1872 is tested in a feedback loop that servos VFB to the output of the error amplifier.
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LTC1872 TYPICAL PERFORMANCE CHARACTERISTICS
Reference Voltage vs Temperature
825 820 815 VIN = 4.2V 10 8
NORMALIZED FREQUENCY (%)
VFB VOLTAGE (mV)
810 805 800 795 790 785 780 775 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
1872 G01
4 2 0 -2 -4 -6 -8 -10 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
1872 G02
UVLO TRIP VOLTAGE (V)
Maximum Current Sense Trip Voltage vs Duty Cycle
130 120
VIN - VSENSE - (mV)
ITH/RUN VOLTAGE (mV)
110 100 90 80 70 60 50 20 30 40 50 60 70 80 DUTY CYCLE (%) 90 100
PIN FUNCTIONS
ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V causes the device to be shut down. In shutdown all functions are disabled and the NGATE pin is held low. GND (Pin 2): Ground Pin. VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output. SENSE - (Pin 4): The Negative Input to the Current Comparator. VIN (Pin 5): Supply Pin. Must be closely decoupled to GND Pin 2. NGATE (Pin 6): Gate Drive for the External N-Channel MOSFET. This pin swings from 0V to VIN.
UW
Normalized Oscillator Frequency vs Temperature
VIN = 4.2V 2.24 2.20 2.16 2.12 2.08 2.04 2.00 1.96 1.92 1.88
Undervoltage Lockout Trip Voltage vs Temperature
VIN FALLING
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1.84 -55 -35 -15
5 25 45 65 85 105 125 TEMPERATURE (C)
1872 G03
Shutdown Threshold vs Temperature
600 560 520 480 440 400 360 320 280 240 200 -55 -35 -15 5 25 45 65 85 105 125 TEMPERATURE (C)
1872 G05
VIN = 4.2V TA = 25C
VIN = 4.2V
1872 G04
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LTC1872 FUNCTIONAL DIAGRA
VIN 5 SENSE - 4
+
ICMP
-
RS1 OSC SLOPE COMP R Q S SWITCHING LOGIC AND BLANKING CIRCUIT VIN NGATE 6
FREQ FOLDBACK
VIN EAMP 0.5A
+
VIN 0.3V
-
VOLTAGE REFERENCE GND 2 UNDERVOLTAGE LOCKOUT VREF 0.8V
OPERATIO
(Refer to Functional Diagram)
Main Control Loop The LTC1872 is a constant frequency current mode switching regulator. During normal operation, the external N-channel power MOSFET is turned on each cycle when the oscillator sets the RS latch (RS1) and turned off when the current comparator (ICMP) resets the latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier EAMP. An external resistive divider connected between VOUT and ground allows the EAMP to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB
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-
+
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0.3V
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+ -
BURST CMP SLEEP
OVP
+ -
VREF + 60mV
0.15V
+ -
VREF 0.8V VFB 3 VIN
1 ITH/RUN
0.35V
+
SHDN CMP SHDN UV
-
1.2V
1872FD
relative to the 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the ITH/RUN pin low. Releasing ITH/RUN allows an internal 0.5A current source to charge up the external compensation network. When the ITH/RUN pin reaches 0.35V, the main control loop is enabled with the ITH/RUN voltage then pulled up to its zero current level of approximately 0.7V. As the external compensation network continues to charge up, the corresponding output current trip level follows, allowing normal operation.
LTC1872
OPERATIO
Comparator OVP guards against transient overshoots > 7.5% by turning off the external N-channel power MOSFET and keeping it off until the fault is removed. Burst Mode Operation The LTC1872 enters Burst Mode operation at low load currents. In this mode, the peak current of the inductor is set as if VITH/RUN = 1V (at low duty cycles) even though the voltage at the ITH/RUN pin is at a lower value. If the inductor's average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the ITH/RUN voltage goes below 0.85V, the sleep signal goes high, turning off the external MOSFET. The sleep signal goes low when the ITH/RUN voltage goes above 0.925V and the LTC1872 resumes normal operation. The next oscillator cycle will turn the external MOSFET on and the switching cycle repeats. Undervoltage Lockout To prevent operation of the N-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the LTC1872. When the input supply voltage drops below approximately 2.0V, the N-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes.
SF = IOUT/IOUT(MAX) (%)
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(Refer to Functional Diagram)
Overvoltage Protection The overvoltage comparator in the LTC1872 will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV. Slope Compensation and Inductor's Peak Current The inductor's peak current is determined by:
IPK = VITH - 0.7 10 (RSENSE )
when the LTC1872 is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 2. Short-Circuit Protection Since the power switch in a boost converter is not in series with the power path from input to load, turning off the switch provides no protection from a short-circuit at the output. External means such as a fuse in series with the boost inductor must be employed to handle this fault condition.
110 100 90 80 70 60 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
1872 F02
IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE VIN = 4.2V
Figure 2. Maximum Output Current vs Duty Cycle
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LTC1872
APPLICATIONS INFORMATION
The basic LTC1872 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the power MOSFET and the output diode D1 is selected followed by CIN (= C1)and COUT(= C2). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor's peak current. The output current the LTC1872 can provide is given by:
0.12 I VIN IOUT = - RIPPLE RSENSE 2 VOUT + VD
where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section) and VD is the forward drop of the output diode at the full rated output current. A reasonable starting point for setting ripple current is:
IRIPPLE = (O.4)(IOUT )
VOUT + VD VIN
Rearranging the above equation, it becomes:
RSENSE = 1 VIN (10)( IOUT) VOUT + VD
for Duty Cycle < 40%
However, for operation that is above 40% duty cycle, slope compensation's effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using Figure 2, the value of RSENSE is:
RSENSE =
VIN (10)(IOUT)(100) VOUT + VD SF
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Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VOUT. The inductor's peak-to-peak ripple current is given by:
IRIPPLE =
VIN VOUT + VD - VIN f (L) VOUT + VD
where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is:
VIN IRIPPLE = 0.4 IOUT(MAX ) VOUT + VD
(
)
In Burst Mode operation, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peak-to-peak ripple current must not exceed:
IRIPPLE 0.03 RSENSE
This implies a minimum inductance of: LMIN = VOUT + VD - VIN VIN 0.03 VOUT + VD f RSENSE
A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods.
LTC1872
APPLICATIONS INFORMATION
Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool M(R) cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool M. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount that do not increase the height significantly are available. Power MOSFET Selection An external N-channel power MOSFET must be selected for use with the LTC1872. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the "on" resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. Since the LTC1872 is designed for operation down to low input voltages, a sublogic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1872 is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation given by: R DS(ON)
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(DC)IIN2(1+ p)
PP
where PP is the allowable power dissipation and p is the temperature dependency of RDS(ON). (1 + p) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but p = 0.005/C can be used as an approximation for low voltage MOSFETs. DC is the maximum operating duty cycle of the LTC1872. Output Diode Selection Under normal load conditions, the average current conducted by the diode in a boost converter is equal to the output load current: ID(avg) = IOUT It is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. A fast switching diode must also be used to optimize efficiency. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection To prevent large input voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current for a boost converter is approximately equal to: CIN Required IRMS (0.3)IRIPPLE where IRIPPLE is as defined in the Inductor Value Calculation section. Note that capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or
Kool M is a registered trademark of Magnetics, Inc.
7
LTC1872
APPLICATIONS INFORMATION
height requirements in the design. Due to the high operating frequency of the LTC1872, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (VOUT) is approximated by: V +V I VOUT IO * OUT D + RIPPLE * VIN 2 ESR2 1 + 2 fCOUT
1 22
where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. The output capacitor RMS current is approximately equal to:
IPK * DC - DC2
NORMALIZED VOLTAGE (%)
where IPK is the peak inductor current and DC is the switch duty cycle. When using electrolytic output capacitors, if the ripple and ESR requirements are met, there is likely to be far more capacitance than required. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX
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TPS, AVX TPSV and KEMET T510 series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Panasonic SP. Low Supply Operation Although the LTC1872 can function down to approximately 2.0V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 3 is the effect of VIN on VREF as VIN goes below 2.3V. Setting Output Voltage The LTC1872 develops a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by: R2 VOUT = 0.8 V 1 + R1
105 100 95 90 85 80 75 2.0 VREF
VITH
2.2
2.4 2.6 2.8 INPUT VOLTAGE (V)
3.0
1872 F03
Figure 3. Line Regulation of VREF and VITH
VOUT LTC1872 3 VFB 100pF R2
R1
1872 F04
Figure 4. Setting Output Voltage
LTC1872
APPLICATIONS INFORMATION
For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, a 100pF capacitor is suggested across R1 located close to LTC1872. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (1 + 2 + 3 + ...) where 1, 2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1872 circuits: 1) LTC1872 DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the contoller's DC supply current. In continuous mode, IGATECHG = f(Qp). 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current sense resistor. The MOSFET RDS(ON) multiplied by duty cycle times the average output current squared can be summed with I2R losses in the inductor ESR in series with the current sense resistor. 4. The output diode is a major source of power loss at high currents. The diode loss is calculated by multiplying the forward voltage by the load current. 5. Transition losses apply to the external MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2(VIN)2IIN(MAX)CRSS(f) Other losses, including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss.
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LTC1872
APPLICATIONS INFORMATION
PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1872. These items are illustrated graphically in the layout diagram in Figure 5. Check the following in your layout: 1. Is the Schottky diode closely connected between the output capacitor and the drain of the external MOSFET? 2. Does the (+) plate of CIN connect to the sense resistor as closely as possible? This capacitor provides AC current to the inductor. 3. Is the input decoupling capacitor (0.1F) connected closely between VIN (Pin 5) and ground (Pin 2)? 4. Connect the end of RSENSE as close to VIN (Pin 5) as possible. The VIN pin is the SENSE + of the current comparator. 5. Is the trace from SENSE - (Pin 4) to the Sense resistor kept short? Does the trace connect close to RSENSE? 6. Keep the switching node NGATE away from sensitive small signal nodes. 7. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. The 100pF capacitor should be as close as possible to the LTC1872.
1
ITH/RUN NGATE LTC1872 GND VIN SENSE -
RITH
2
3 CITH
VFB
C1
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)
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VIN
+
5 RS 0.1F 4 L1 CIN M1 D1 VOUT COUT R2
+
R1
1872 F05
LTC1872
TYPICAL APPLICATIO
10k 220pF
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSE476M016R0150 D1: MOTOROLA MBR0520LT1
PACKAGE DESCRIPTION
0.35 - 0.55 (0.014 - 0.022)
NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DIMENSIONS ARE INCLUSIVE OF PLATING 3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 4. MOLD FLASH SHALL NOT EXCEED 0.254mm 5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
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LTC1872 3.3V to 12V/100mA Boost Converter
R1 0.15 1 ITH/RUN LTC1872 2 3 GND VFB SENSE - NGATE 4 6 M1 D1 VIN 5 L1 22H C1 10F 10V VIN 3.3V
+
C2 47F 16V
VOUT 12V 100mA 1.1M
L1: COILTRONICS UP1B-220 M1: Si9804DV R1: DALE 0.25W
78.7k
1872 TA02
Dimensions in inches (millimeters) unless otherwise noted. S6 Package 6-Lead Plastic SOT-23
(LTC DWG # 05-08-1634)
2.80 - 3.00 (0.110 - 0.118) (NOTE 3)
2.6 - 3.0 (0.110 - 0.118) 1.50 - 1.75 (0.059 - 0.069)
1.90 (0.074) REF 0.00 - 0.15 (0.00 - 0.006)
0.95 (0.037) REF
0.90 - 1.45 (0.035 - 0.057)
0.09 - 0.20 (0.004 - 0.008) (NOTE 2)
0.35 - 0.50 0.90 - 1.30 (0.014 - 0.020) (0.035 - 0.051) SIX PLACES (NOTE 2) S6 SOT-23 0898
11
LTC1872
TYPICAL APPLICATIO
1 0.1F CERAMIC
+
C1 100F 10V
ITH/RUN LTC1872 GND VFB
VIN SENSE - NGATE
5
L1 4.7H
10k 220pF
2 3
4 6 M1 D1 332k
VIN -2.5V C1, C2: AVX TPSE107M010R0100 D1: MOTOROLA MBR2045CT L1: COILTRONICS UP2B-4R7 M1: Si9804DV R1: DALE 0.25W U1: PANASONIC 2SB709A x2
RELATED PARTS
PART NUMBER LT1304 LT1500/LT1501 LT1572 LT1610 LT1613 LT1680 LTC1624 LT1615 DESCRIPTION Micropower DC/DC Converter with Low-Battery Detector Adaptive Frequency Current Mode Switching Regulators 100kHz, 1.25A Switching Regulator with Catch Diode 1.7MHz, Single Cell Micropower DC/DC Converter 1.4MHz, Single Cell DC/DC Converter in 5-Lead SOT-23 High Power DC/DC Step-Up Controller High Efficiency SO-8 N-Channel Switching Regulator Controller Micropower Step-Up DC/DC Converter in SOT-23 COMMENTS 120A Quiescent Current, 1.5V VIN 8V 700mA Internal Power Switch, 500kHz, 1.8V VIN 15V 16-Pin SO Package, 3V VIN 30V 30A Quiescent Current, VIN Down to 1V Internally Compensated, VIN Down to 1V Operation Up to 60V, Fixed Frequency Current Mode 8-Pin N-Channel Drive, 3.5V VIN 36V 20A Quiescent Current, VIN Down to 1V
1872i LT/TP 0700 4K * PRINTED IN USA
12
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com
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LTC1872 - 2.5V to 3.3V/0.5A Boost Converter
R1 0.034 C2 2x 100F 10V VOUT 3.3V 0.5A U1 100pF 80.6k
1872 TA03
180k
(c) LINEAR TECHNOLOGY CORPORATION 2000

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